In the last decades, the demand for a low-cost, day-night, all-weather spaceborne imaging capability pushes to further development of synthetic aperture radar (SAR) systems, as satellite payload mainly for Earth observation (EO) applications. SAR payloads exhibit a very good spatial resolution and operation in several RF bands, such as S, C, X, and Ka [1]. After the success of SAR payloads on board of satellites having a mass of some tons, recently, the interest has also been focused on smaller satellites, with mass in the range 100–500 kg, and their constellations [2]. Several SAR payloads for space missions based on standard electronic components have been launched [3].

A conventional SAR payload includes a Linearly Chirped Microwave Waveform Generator, frequency converters, analog-to-digital converters, and a beamforming network. In the transmission chain, the chirped microwave signal, is in-phase/quadrature modulated, up-converted, and amplitude-limited. The resulting signal drives a gain-controlled amplifier that feeds the beamforming network and the antenna system. In the reception chain, the echo signal, received by the antenna, is phase- and amplitude-handled by the beamforming/beamsteering network. The output signal is amplified and post-processed by down-conversion, I/Q demodulation, and baseband filtration. At the end, the target image is a result of A/D data conversion [4].

Since 2000s, photonic solutions for SAR payloads have been investigated, by exploiting the photonics benefits [5], in terms of high-frequency operation, high immunity to external perturbations, lightness, compactness, high immunity to electromagnetic interference, and power consumption.

Encouraging results on photonic SAR key building blocks have been reported in the literature [6–8], paving the way to light, compact, and high performing photonic SAR payloads for next-generation space missions.

In the transmission section of a SAR payload, the LCMWG plays a crucial role to achieve large detection range and high-range resolution, at the same time. The LCMWs with carrier frequency and bandwidth larger than tens of gigahertz and TBWP ranging from 10² to 10³ are desired to achieve an efficient microwave pulse compression at millimeter wave band [9]. In fact, since the range resolution of a pulsed radar system is limited by the bandwidth of the transmitted pulse, a wide bandwidth can be achieved by using a long pulse with linear frequency modulation also preserving the radiometric resolution. Large TBWPs entail high-range resolution.

Although recently some interesting electronic LCMWGs [10] have been proposed, the intrinsically limited speed and bandwidth of the electronics remain the main bottlenecks, with a resulting central frequency and bandwidth <10 GHz, that clash with the next-generation SAR requirements. These constraints could be overcome by the microwave photonics technology [11], whose intrinsic large operating bandwidth has the potential of achieving a microwave-compressed pulse at giggahertz frequency with large TBWP.

Photonic approaches for LCMW generation proposed in the literature are mainly based on temporal pulse shaping, wavelength/frequency/space-to-time mapping technique, optically injected lasers, interferometers, and optoelectronic oscillators [12–27]. The temporal pulse shaping technique is based on the amplitude modulation of an ultrashort optical signal and the processing by using conjugated or mismatched chromatic dispersion elements. TBWPs in the order of tens have been demonstrated, limited by the intensity modulator operation [12, 13].

In the wavelength/frequency/space-to-time mapping technique, the chirped waveform is performed by the optical-to-electrical conversion of the shaping pulse, with resulting TBWPs > 100 at frequencies >10 GHz [14–18].

To improve the TBWP, the injected laser technique has been proposed but at the expense of the system compactness. It exploits the period-one dynamics of an optically injected modulated laser [19] or the dual-mode state of a single monolithically integrated amplified feedback laser [20]. Ultrahigh TBWP (> 10⁵) has been demonstrated but with a significant increase in phase noise. Also, optical interferometric configurations have been investigated, by using a Mach–Zehnder interferometer [21] or Sagnac loop [22], targeting to achieve high performance (TBWP > 10³) preserving the system compactness. However, these systems suffer small reconfigurability of the operating frequency and large phase noise induced by using non-coherent laser sources.

A technique based on optoelectronic oscillators (OEOs) was also proposed in [23], where the OEO includes a phase modulator and a phase-shifted fiber Bragg grating, jointly operating as a tunable notch microwave filter that is used to phase-modulate the signal at the grating. Tunability is obtained by tuning the wavelength of the optical carrier.

To further improve the TBWP, in addition to a frequency-tunable OEO, a recirculating phase modulation loop (RPML) was proposed [24]. The frequency-tunable microwave signal generated by the OEO and a chirp-free pulse are sent to the RPML, where the chirp-free pulse is phase modulated by a parabolic waveform to obtain a linearly chirped optical waveform. TBWP can also be enhanced by properly engineering the driving voltage of the modulators, for example, by using pseudo-random sequence, splitting parabolic waveform, or both. In particular, the use of a mode-locked laser and a phase encoding with pseudo-random sequence has demonstrated a TBWP up to 80,000 [25], but the system becomes more complicated and not reconfigurable. By driving a polarization modulator with M steps—parabolic waveform, a TBWP improvement by M/2 times with respect to an unsplit parabolic driving signal has been proved [26]. The merging of the two approaches has been investigated in [27], where the cascade of the dual-polarization Mach–Zehnder modulator and a polarization modulator, driven by a parabolic microwave waveform with pseudo-random sequence, reveals a TBWP of 20,160 with a central frequency of 40 GHz. This result is strictly related to the shape of the driven voltage and to the sampling rate of the electrical waveform generator but whose feasibility was not demonstrated. Moreover, it is worth noticing that in general, the use of discrete optical components affects the volume of the whole system, in contrast with the compactness requirements for light-scale next-generation SAR payloads for small satellites. However, monolithic [28] and heterogeneous [29] approaches allow the development of optoelectronic active and passive devices, performing high-efficiency functionalities within a single chip, with a resulting powerful and ultracompact system.

In this article, we propose a compact LCMWG architecture, based on RPML and OEO sections, whose key building blocks are a phase modulator (PM) and a high Q-factor ring resonator (RR), respectively. By properly driving the phase modulators in the RPML, we have calculated a TBWP of 1,072 at baseband. In the OEO section, a sinusoidal signal with ultralow phase noise (<100 dBc/Hz) at 40 GHz has been achieved by using a very high Q-factor, of the order of 10¹⁰, Si₃N₄-based ring resonator with a 1D superimposed grating. By beating the RPML and OEO output signals, an LCMW signal with TBWP of 1,078 and pulse compression rate (PCR) of 814 have been calculated. These ultrahigh TBWP and PCR, with large signal purity, result from an innovative design of the OEO and RPML sections.

The proposed system is a compact, reconfigurable, and potentially integrable within a single heterogeneous chip, paving the way to scalable and high-performing system fulfilling the requirements of next-generation SAR payloads.

The article is organized as follows: in Section 2, the principle of operation of a photonic LCMW generator is described. In Section 3, the design of the LCMW, including RPML and OEO, is carried out. In Section 4, the overall performance of the generator is discussed. Finally, the conclusions are in Section 5.

The proposed architecture consists of two sections, a tunable OEO and an RPML, as shown in Figure 1, conceptually similar to those reported in Ref [24]. Some photonic devices included in this scheme have been designed, and specifications of others have been defined to ensure the potential heterogeneous integration of the whole system on a single chip, following the technique reported in [29]. RF components of the OEO can be integrated on an MMIC, which should be co-packaged with the photonic integrated circuit (PIC), in order to make the LCMW a system-in-package.

The beam generated by an InP continuous-wave laser source, including a Bragg grating, with an output power of 20 mW, a linewidth ≤2 kHz and a relative intensity noise (RIN) of −140 dB/Hz [30], is routed in two branches by using a 3-dB power splitter (PS₁) [31].

One signal is intensity modulated by an InP-based Mach–Zehnder modulator (MZM) [32], electrically driven by a square wave voltage with falling edge at time T ₀ , that represents the target temporal duration of the chirped signal. The MZM output enters the RPML section, where the signal is phase modulated by a traveling wave InP phase modulator (PM₁) with a bandwidth up to 80 GHz, V _π = 1.5 V, and insertion loss (IL) equal to 8.5 dB [33], driven by an engineered electric parabolic waveform, amplified by a semiconductor optical amplifier (SOA₁) [34] and temporally delayed by an optical delay line (DL). When the recirculating signal carries out N roundtrips within the RPML, reaching the desired chirp rate, it is routed out by using a silicon MZI switch [35], electrically driven by a square wave voltage with rise edge time at N × T ₀ + N × τ, where τ is the time delay of each round trip within the RPML.

The other PS₁ signal is phase modulated by a PM and then routed out by a PS₂. The signal within the OEO is amplified by a SOA₃ and then O/E transduced by a photodiode (PD₁). The electrical signal is RF filtered and amplified, phase shifted [36], and then, fed back in the PM, generating the oscillating signal.

The output of OEO and RPML, properly filtered by a compact high-extinction ratio (ER) filter, and amplified by SOA₂, beats at a photodiode (PD₂), to generate the LCMW signal. Both PDs operate at 1.55 µm with a responsivity of 0.7 A/W, a dark current of 150 nA, a total resistance of 100 Ω, and a bandwidth >60 GHz [37].

When the MZM and SW are in “on” state, the lightwave E _RP (t) at the output of the RMPL section can be expressed as follows [38]: E RP ( t ) = E 0 , RP ⋅ exp [ j ( 2 π f 0 t  +  N π V pp V π c ( t ) ) ] , (1) where E _0,RP is the amplitude of E _RP , f ₀ is the operating frequency of the light source (≈193 THz), N is the number of round-trips in the RMPL section, V _pp is the peak-to-peak voltage of the parabolic waveform, and V _π is the half-wave voltage of the PM.

s(t) is the driven voltage of the MZM, whose output E _MZM can be expressed as follows [39]: E MZM = E CW 2 ( e jπs ( t ) 2 V π , MZM + e − jπs ( t ) 2 V π , MZM ) = E CW cos ( π 2 s ( t ) V π , MZM ) , (2) s ( t )   =   {   0 t  ≤   T 0 V π , MZM for other t values , (3) where V _π,MZM is the half-wave voltage of the MZM and E _CW is the laser output beam. When s(t)= V _π, the output signal is null (“off” state), while, s(t) = 0, E _MZM = E _CW (“on” state). s w(t) is the driven voltage of the switch, SW, sw ( t )   =   {   V π , SW t  ≥ N  ×   T 0 + N  ×  τ 0 for other t values     , (4) where V _π,MZM is the half-wave voltage of the MZI-based switch. When sw(t)= V _π,SW , the output signal is routed out of RPML (“on” state).

c(t) is the split parabolic waveform that drives the PM, c ( t ) = Ω +   K  ·  M 2 { ( t − T 0 / 2 ) 2 − ( t n ( 1 ) − T 0 / 2 ) 2 t n ( 0 )   ≤  t  ≤   t n ( 1 ) ( t − T 0 / 2 ) 2 − ( t n ( 2 ) − T 0 / 2 ) 2 ⋮ t n ( 1 )   ≤  t  ≤   t n ( 2 ) ⋮ ( t − T 0 / 2 ) 2 − ( t n ( M / 2 ) − T 0 / 2 ) 2 ( t − T 0 / 2 ) 2 − ( t p ( 0 ) − T 0 / 2 ) 2 ( t − T 0 / 2 ) 2 − ( t p ( 1 ) − T 0 / 2 ) 2 ⋮ ( t − T 0 / 2 ) 2 − ( t p ( M / 2 − 1 ) − T 0 / 2 ) 2 t n ( M / 2 − 1 )   ≤  t  ≤   t n ( M / 2 ) t p ( 0 )   ≤  t  ≤   t p ( 1 ) t p ( 1 )   ≤  t  ≤   t p ( 1 ) ⋮ t p ( M / 2 − 1 )   ≤  t  ≤   t p ( M / 2 ) , (5) where M is the number of splitting, T ₀ is temporal duration of the signal, and Ω = (0,—0.5) for non-return-to-zero (NRZ) or return-to-zero (RZ) c(t) signal, and  K =   4 T   0   2   ,     t n ( i ) = T 0 2 − M / 2 − i K ·  M / 2   , and t p ( i ) = T 0 2 + i K ·  M / 2   .

The split parabolic waveform c(t) results by the split into M pieces, ranging from t _p (i-1) to t _p (i) or t _n (i-1) to t _n (i), of the parabolic signal and the amplitude of each piece is set to 1. When the MZM and/or SW are in “off” state, E _RP (t) = 0.

Since the OEO output presents the carrier at f ₀ and optical sidebands at f ₀ ± f _OEO , f0 ± 2 × f _OEO , f ₀ ± 3 × f _OEO ,…, the lightwave E _OEO (t) at the output of the OEO section can be expressed as follows: E OEO ( t ) = E 0 , OEO ⋅ exp [ j ( 2 π ( f 0   −   f OEO ) t ) ] , (6) where E _0,OEO is the amplitude of E _OEO , f _OEO is the oscillating frequency. In the Ka-band, it results f _OEO ≈ 193.04 THz at 40 GHz.

The LCMW is generated by beating E _RP (t) and E _OEO (t) at the PC. By neglecting the DC component, the PD output i(t) can be expressed as follows:   i ( t ) ∝   [ E RP ( t ) + E OEO ( t ) ] 2 = { cos [ 2 π f OEO t 1 + Nπ V pp V π ( Ω + MK 2 ( ( t 1 − T 0 2 ) 2 −   a i ) ) ] t 1    ≤   T 0 0 for other  t 1  values , (7) where t 1 = t − ( N  ×   T 0 + N  ×  τ ) , a _i is extracted by Eq. 5 ( a i =   ( t n ( i ) − T 0 / 2 ) 2 for t n ( 0 )   ≤    t 1 ≤   t n ( M / 2 ) and 1   ≤  i  ≤  M / 2   , a i =   ( t p ( i ) − T 0 / 2 ) 2   for t p ( 0 )   ≤   t 1   ≤   t p ( M / 2 ) , and 0   ≤  i  ≤  M / 2   − 1 ) .

The TBWP has been calculated after determining the instantaneous frequencies (f _ist,max , f _ist,min ) from Eq. 7: T B W P   = T 0 | f i s t , max − f i s t , min | =     2   ⋅ M   ⋅ N   ⋅ V p p V π . (8)

The values f _ist,max and f _ist,min represent the maximum and minimum instantaneous frequencies of the signal i(t), respectively, calculated as f i s t =   1 / 2 π   ∂ ∠ i ( t ) / ∂ t .

The TBWP trend vs. V _pp , M, and N is reported in Figure 2, by assuming a signal duration T ₀ = 15 ns. As expected, the TBWP increases as M, N, or V _pp increases, while the number of roundtrips N affects the response time of the LCMW generator. TBWP should be designed as the best compromise between the response time and the target TBWP, also in accordance with the values M and V _pp (Eq. 8). In order to keep the power consumption low, an upper V _pp value of 5 V has been considered, while the M upper limit should be set in accordance with the splitting capability of the electrical waveform generator.

To ensure the feasibility of the desired split parabolic waveform, an electronic circuit has been designed. The splitting parabolic waveform is generated by the circuit in Figure 3A, where the desired signal results from the difference between the filtered parabolic signal s ₁ (t), with time duration T ₀ , and an engineered step voltage signal s ₂ (t), as described in the following.

RZ/NRZ (Figures 3B,C) parabolic signal (s₁(t) in Figure 3A), generated by a parabolic waveform generator (PARABOLIC WF GEN), based on a sawtooth generator and synch pulse generator both with period T ₀ [40], is filtered by a single half-wave rectifier, implemented with a low threshold voltage diode (FIL) [41], and then, it is power split by a 3dB divider (SP), generating two replicas.

One replica feeds an (M/2–1) × 1 array of voltage comparators (V-COMP) [42], where the voltage references V _ref have been set in accordance with V _pp and M. As example, for an RZ chirped signal, V _pp = 5 V and M = 10, 4 voltage comparators are needed, with V _ref,1 = 4 V, V _ref,2 = 3 V, V _ref,3 = 2 V, and V _ref,4 = 1 V. The voltage comparators weighted outputs are added together in a high-speed OPAMP [43], generating the signal s ₂ (t), which shows a step behavior.

The parabolic split signal results from the differential sum, carried out by an OPAMP, of the two electrical signals, s ₁ (t) and s ₂ (t). The amplitude of the resulting electrical signal can be tailored by properly engineering the impedances Z ₁ , Z ₂ , Z ₃ , and Z ₄ .

In terms of time resolution, the fall/rise edge time of voltage comparators represents the main limitation. Since the V-COMP shows a first-order behavior with 20–80% fall/rise edge time values <40 ps[42], the whole comparator input/output dynamic (0—≈ 99.9%) lasts more than 170 ps. To achieve a split piece larger than 170 ps, M could range up to 40. As example, a TBWP ≈ 600–1,200 can be calculated from Eq. 8, by considering N = 20, V _pp = 1.5 V, V _π = 1.8 V, and M = 20–40, respectively.

The LCMW signal is generated through a fast PD₂ after combining the RPML and OEO output signals. The RF spectra for M = 40, N = 9 and M = 40, N = 18 are shown in Figures 9A–L, calculated by using the system-level simulator Optisystem®. For OEO based on Si₃N₄—PhCRR, described in Section 3.2, TBWP of 595/1,078 at 40 GHz and RFSRR of 13 dB/11 dB have been calculated for N = 9/18, respectively. The autocorrelation function of the LCMW shows an ultra-large PCR of 346/814 and PSR = 8.85/6.79 dB, for N = 9/18, respectively. The TBWPs are compliant with the predicted ones according to Eq. 8.

In both cases, the bandwidth of the LCMW is of the order or even larger (40 GHz, for N = 9, and 72 GHz, for N = 18) than the central frequency (40 GHz), with a consequent partial overlapping between the signal at 40 GHz and tones at 0 and 80 GHz.

Moreover, RFSRR gets worse with respect to the RPML values; it also decreases with increasing TBWP.

About the frequency sweeping linearity, as shown by Figures 9E,I, the linear fitting of the instantaneous frequencies over the time shows R ² equal to 0.9982 and 0.9855 for N = 9 and N = 18, respectively. This statistical measure demonstrates the linearity of the achieved compressed signal, which is useful for practical SAR systems.

To point out the crucial role of high-Q ring resonators in the whole system, radiation-hard InP RRs, with a radius equal to 13 mm, Q-factor of the order of 10⁶, and ER around 6 dB [58, 59], and bared Si₃N₄ ring resonators, with a radius equal to 6 mm, Q-factor = 1.02 × 10⁶, and ER = 13.1 dB have been investigated.

For both configurations, the achieved OEO phase noise (≈−50 dB for InP-RR and −47 dB for Si₃N₄—RR) involves a noisy relevant LCMW (ripple >10 dB), with an RFSSR less than 5 dB. Hence, low-phase noise OEO and high Q-factor RR, as PhCRR, are needed to enhance the LCMWG performance, in terms of TBWP, PCR, and PSR.

Furthermore, in the SAR context, the frequency tunability is very important to make the system immune to jamming, atmospheric, and mutual interference. The proposed system results are also suitable to fit the requirements of SAR adjustments and optimization after the launch. In particular, the central frequency of the chirped signal can be tuned within a frequency range of tGHz, by simply varying the input laser frequency and thermo-optically/electro-optically tuning the RR resonance frequency or both. The system reconfigurability is limited by the bandwidth of the RF filter (2 GHz). The RF filter could be omitted, at the expense of a larger phase noise (−88 dBc/Hz), or several countermeasures can be adopted, that is, by using a matrix of RF filters. Overcoming the RF filter limited tunability, the upper limit of the system reconfigurability is strictly dependent on the bandwidth of the PM (<80 GHz) and PD (<60 GHz). Therefore, according to the well-known insensitivity of the photonic devices to the operating frequency, TBWPs > 1,000 with a central frequency up to 60 GHz are envisaged. To further improve the reconfigurability, additional investigation is needed in order to make the proposed generator suitable for applications that require an ultra-large (>100 GHz) frequency tunability.

We have proposed a compact integrable LCMW generator, based on an RPML and OEO sections, to be used as a key building block in photonic SAR systems. The design of the whole system has been carried out aiming to achieve a TBWP ≈ 10²–10³, to ensure large range resolution and large RFSRR, to preserve the output signal purity. To fulfill the TBWP requirements, LCMW has been tailored by properly engineering the number of round trips N within the RPML section and the parabolic waveform, with M split pieces, which drives the PM₁. To improve the RFSRR and the signal purity, a Si₃N₄ ring resonator with a photonic crystal has been considered, reporting a Q-factor of 2 × 10¹⁰ with ER = 8 dB. OEO phase noises of −100 dBc/Hz have been calculated. In the best-case scenario, TBWP ≈ 1,078, with ultra-large PCR = 814, and RFSRR ≈ 11 dB have been calculated at 40 GHz, by considering V _pp = 1.1 V, M = 40, and N = 18. Different solutions of a ring resonator with different Q-factors, as the hearth of the OEO, have been also investigated to assess the Q-factor impact on the LCMW performance. All the generator sections have been designed as compliant to the OEO heterogeneous integration on a silicon substrate by a CMOS-compatible technological process, forming a system-in-package. The proposed device overcomes the main limitations of similar architectures, which mainly regard the use of discrete components, with a lack of compactness. The reported performance (TBWP > 10³ in Ka band, within a compact footprint) makes the proposed system suitable for next-generation SAR system with large range resolution request.